More and more complex functions are implemented in modern integrated circuit devices that need to be very efficient for keeping as low as possible their operating temperature. Class D audio amplifiers, which are PWM amplifiers, are particularly suitable for satisfying stringent requirements on power dissipation.
The functioning of a class D amplifier is based on the modulation of a high frequency carrier by the signal to be amplified, and on the successive reconstruction of the amplified signal. In particular, in a pulse width modulation (PWM) mode the duty-cycle of a square wave signal, modulated at a constant frequency much greater than the frequency band of the signal to be amplified, is varied depending on the instantaneous value of the input voltage. Ideally, the output signal of a class D amplifier is a square wave, with a fixed period and amplitude adequate to obtain the desired maximum output power. The sole parameter that varies is the duty cycle, that is, the ratio between the time during which the output remains high and the switching period.
Information of the signal to be amplified is contained in the mean value of the output value. This is normally recovered through a low-pass filter with passive L-C components. The reconstruction filter LC makes the output current remain substantially stable around its mean value.
In class A, AB, and B amplifiers, the active element is biased such to dissipate a large power for the entire conduction phase. In class D amplifiers, this does not happen because the transistors ideally always work under conditions of null power consumption: VDS=0 when they are in a conduction state (ON); and ID=0 when they are in a non-conductive state (OFF).
Theoretically, the efficiency is 100%, that is, all the power absorbed from the power supply is delivered to the load. FIG. 1 shows the equations for calculating the power and a basic control circuit of the final stage of a PWM half-bridge.
According to a standard technique for driving the power transistors of a half-bridge stage, when the transistor is in a forward conduction state (that is, when out of the dead times and the output is low), the gate-source voltage VGS is kept at a value, for example 6V, equal to that for which the on-state drain-source resistance RDSon is minimum while keeping the integrity of the gate oxide. Thus, the drain current ID can circulate in both directions depending on the sign of the instantaneous value of Iout.
A control circuit that implements this technique is depicted in FIG. 2. The block SYNC indicates a circuit for commanding the switches S1 and S2, which for example may be the circuit disclosed in U.S. Pat. No. 6,288,605 that is assigned to the current assignee of the present invention.
During dead times, the gate-source voltages VGS of the transistors are normally brought to 0V, the transistors are in an OFF state and one of the two free-wheeling diodes is in a conduction state.
A drawback of this type of amplifier is a relatively high electromagnetic emission (EMI) that disturbs other circuits connected to the same supply lines of the amplifier. This problem is particularly felt in devices that include radio receivers, such as for example car radios, GPS, cellular phones and the like because the emissions generated by the amplifier may interfere with signals coming from antennas.
Electromagnetic interferences (EMI) are due to abrupt variations of the voltage or of the current and to relatively high values of the time derivative of voltages or currents in the power stage. Because of the functioning mode of class D amplifiers, it is difficult to avoid abrupt variations of the output voltage, thus they will always generate electromagnetic disturbances or EMI. Nonetheless, it is well known that there are other causes of electromagnetic emissions in class D amplifiers that may be effectively limited.
The free-wheeling diode that is normally present in every output power stage that drives an inductive load, causes current peaks that may also generate electromagnetic interferences. Typically, in a half-bridge stage, the free-wheeling or recirculation diode is connected in parallel to the respective power transistors T1 and T2, and is either integrated with the power MOS device or it is intrinsic to the integrated structure of the power MOS device, as shown in FIG. 3.
As it is well known and explained in the article by M. Berkhout, “An Integrated 300-W Class-D Audio Amplifier”, IEEE Journal of Solid-State Circuits, Vol. 48, No. 9, July 3003, the free-wheeling diode needs a non-null time for passing from the conduction state to the OFF state generates EMI.
The presence of a dead time, that is, a time interval during which both switches are turned off and the output current is kept practically constant by the LC filter, causes the turning on of one of the free-wheeling diodes during the dead time.
To understand how EMI is generated when a free-wheeling diode switches from a conduction state to a cut-off state, let us consider the situation depicted in FIG. 3, where the MOS transistors T1 and T2 are off and the first transistor to turn on is T1. In this situation, the free-wheeling diode in parallel to the MOS transistor T2 is in a conduction state.
The critical phase starts when the MOS transistor T1 enters in a conduction state. When the transistor T1 is on, the output voltage Vout needs to reach the positive supply voltage. Even if relatively short, the diode D2 needs a certain time for turning off, and during this time the voltage on its nodes is kept practically constant.
In this situation, the MOS transistor T1 works in a saturation zone with a VGS that becomes sufficiently high to circulate a relatively large current (about 15-20 A). This large current lets minority carriers recombine in the diode by turning it off in a very short time (on the order of tenths of ns), and the output voltage Vout reaches the positive supply voltage. Therefore, there is a very short time interval in which the MOS transistor T1 and the free-wheeling diode of the MOS transistor T2 are both in a conduction state. Thus, the power supplies are shorted and a narrow current pulse of very large amplitude circulates through the supply lines.
With the illustrated control technique, a signal having a broad high frequency spectrum is forced through the supplies, and it is capable of disturbing the functioning of electronic circuits connected to the same supply line.
FIG. 4 depicts typical waveforms of the current flowing in the MOS device of the half-bridge, with evident cross-conduction peaks between the positive and negative power supplies, due to the above-illustrated phenomenon.
The output voltage Vout of the switching stage may be approximated with a square wave with infinitely steep edges only at a first level of approximation. It is better modeled with a trapezoidal wave of non-null rise and fall times as illustrated in FIG. 5. This waveform has a spectrum that may be calculated as the product of the spectrum of a rectangular signal of duration Ton with the spectrum of another rectangular signal of duration Trise (rise time).
These spectra are described by sinc functions of different periods. FIG. 6 compares the spectra of an ideal output voltage, wherein a sinc function the zeros are at frequencies very close to each other, are enveloped by the function with zeroes at frequencies more distant from each other, and of the output effective voltage Vout.
By adjusting the slope of the transition edges ON-OFF and OFF-ON, it is possible to obtain spectral zones at pre-established frequencies in which the harmonic components of the output voltage are attenuated.
FIG, 7 depicts a basic scheme of a known PWM driver with control of the slopes of the switching edges of the output voltage. In this case, the power MOS transistors are driven by a buffer of a unitary gain, and the current generators Ip(on) and Ip(off) are connected in input thereto.
The generator Ip(off) is connected when the transistor T1 needs to be turned off. It may be for example realized with a P-channel MOS or a bipolar PNP transistor biased to work as a constant current generator that turns off when the voltage input to the buffer reaches the maximum voltage +Vcc. In this situation, the P-channel MOS transistor or the bipolar PNP transistor does not inject anymore a current through the capacitor Cp.
The generator Ip(on) is connected when the MOS transistor T1 needs to be turned on and it may be realized with an N-channel MOS transistor or a bipolar NPN transistor. As for the generator Ip(off), it does not inject any current through the capacitor Cp when the output voltage reaches the minimum value +Vcc−Vp(on).
The current input to the buffer is negligible, thus the voltages on the capacitors Cp or Cn, feedback from the node on which the output voltage Vout is produced and the inputs of the buffer, may vary with a slope that is a function of the ratio Ip/Cp and In/Cn.
Because the transitions of the output voltage take place at a gate voltage that is practically constant compared to the output voltage swing, the edges of the output voltage should have similar slopes that may be easily controlled by regulating the currents Ip and In.
Moreover, the frequencies at which the zeroes of the envelope are located, depend on the duty-cycle of the output voltage, but only on the slope of the switching edges. Therefore, once the slew-rate with which the output voltage must switch and also the frequency band at which the low emission is fixed, will no longer depend on the value of the input voltage of the amplifier.
Nonetheless, a trade-off is necessary between the location of the zero and the efficiency of the amplifier. The efficiency of a class D amplifier is ideally equal to 100% if power dissipation in the active elements (MOS transistors) of the amplifier is negligible.
This ideal situation would be reached only if switching times of the output and the on resistances of the transistors were both null. As a matter of fact, none of these two parameters can be made absolutely null.
The power consumption of the amplifier varies linearly with the ratio between the switching time Trise and the period T. The longer the Trise, the greater the power dissipation. Thus, it is possible to obtain a substantial reduction of electromagnetic interference at the cost of reducing the efficiency of the class D amplifier.
This approach is applied in a method of controlling the switching of a brushless motor, disclosed in the U.S. Pat. No. 5,191,269. A system of controlling the switching slew-rate of a power stage is disclosed in the U.S. Pat. No. 5,469,096.